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1 MHz to 8 GHz, 60 dB Logarithmic Detector/Controller AD8318
FEATURES
Wide bandwidth: 1 MHz to 8 GHz High accuracy: 1.0 dB over 55 dB range (f < 5.8 GHz) Stability over temperature: 0.5 dB Low noise measurement/controller output (VOUT) Pulse response time 10 ns/12 ns (fall/rise) Integrated temperature sensor Small footprint CSP package Power-down feature: <1.5 mW at 5 V Single-supply operation: 5 V @ 68 mA Fabricated using high speed SiGe process
FUNCTIONAL BLOCK DIAGRAM
VPSI ENBL TADJ VPSO
TEMP
TEMP SENSOR
GAIN BIAS
SLOPE
I
V
VSET
I DET INHI INLO DET DET DET
V
VOUT
CLPF
CMIP
CMOP
Figure 1.
APPLICATIONS
RF transmitter PA setpoint control and level monitoring RSSI measurement in base stations, WLAN, WiMAX, and radar
GENERAL DESCRIPTION
The AD8318 is a demodulating logarithmic amplifier, capable of accurately converting an RF input signal to a corresponding decibelscaled output voltage. It employs the progressive compression technique over a cascaded amplifier chain, each stage of which is equipped with a detector cell. The device is used in measurement or controller mode. The AD8318 maintains accurate log conformance for signals of 1 MHz to 6 GHz and provides useful operation to 8 GHz. The input range is typically 60 dB (re: 50 ) with error less than 1 dB. The AD8318 has a 10 ns response time that enables RF burst detection to beyond 45 MHz. The device provides unprecedented logarithmic intercept stability vs. ambient temperature conditions. A 2 mV/C slope temperature sensor output is also provided for additional system monitoring. A single supply of 5 V is required. Current consumption is typically 68 mA. Power consumption decreases to <1.5 mW when the device is disabled. The AD8318 can be configured to provide a control voltage to a VGA, such as a power amplifier or a measurement output, from Pin VOUT. Because the output can be used for controller applications, wideband noise is minimal. In this mode, the setpoint control voltage is applied to VSET. The feedback loop through an RF amplifier is closed via VOUT, the output of which regulates the amplifier output to a magnitude corresponding to VSET. The AD8318 provides 0 V to 4.9 V output capability at the VOUT pin, suitable for controller applications. As a measurement device, Pin VOUT is externally connected to VSET to produce an output voltage, VOUT, which is a decreasing linear-in-dB function of the RF input signal amplitude. The logarithmic slope is nominally -25 mV/dB, but can be adjusted by scaling the feedback voltage from VOUT to the VSET interface. The intercept is +20 dBm (re: 50 , CW input) using the INHI input. These parameters are very stable against supply and temperature variations. The AD8318 is fabricated on a SiGe bipolar IC process and is available in a 4 mm x 4 mm, 16-lead LFCSP package, for the operating temperature range of -40oC to +85oC.
Rev. A
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c)2006 Analog Devices, Inc. All rights reserved.
04853-001
AD8318 TABLE OF CONTENTS
Features ...................................................................................................1 Applications............................................................................................1 Functional Block Diagram ...................................................................1 General Description ..............................................................................1 Revision History ....................................................................................2 Specifications..........................................................................................3 Absolute Maximum Ratings.................................................................6 ESD Caution.......................................................................................6 Pin Configuration and Function Descriptions..................................7 Typical Performance Characteristics ..................................................8 Theory of Operation .......................................................................... 11 Using the AD8318 .............................................................................. 12 Basic Connections .......................................................................... 12 Enable Interface .............................................................................. 12 Input Signal Coupling.................................................................... 12 Output Interface ............................................................................. 13 Setpoint Interface ........................................................................... 13 Temperature Compensation of Output Voltage..........................13 Temperature Sensor ........................................................................14 Measurement Mode ........................................................................14 Device Calibration and Error Calculation...................................15 Selecting Calibration Points to Improve Accuracy over a Reduced Range ................................................................................16 Variation in Temperature Drift from Device to Device.............17 Temperature Drift at Different Temperatures .............................17 Setting the Output Slope in Measurement Mode .......................17 Response Time Capability .............................................................18 Output Filtering...............................................................................18 Controller Mode..............................................................................19 Characterization Setup and Methods...........................................21 Evaluation Board .................................................................................22 Outline Dimensions............................................................................24 Ordering Guide ...............................................................................24
REVISION HISTORY
1/06--Rev. 0 to Rev. A Changed TADJ Resistor to RTADJ Resistor....................Universal Changes to Applications .................................................................. 1 Changes to Table 1............................................................................ 3 Changes to Figure 5, Figure 6, and Figure 7 Captions................. 8 Changes to Figure 12 Caption......................................................... 9 Changes to Figure 15 Caption......................................................... 9 Changed General Description Heading to Theory of Operation ...................................................................... 11 Changes to Enable Interface Section............................................ 12 Inserted Figure 24........................................................................... 12 Changes to Input Signal Coupling Section ................................. 12 Changes to Measurement Mode Section..................................... 14 Changes to Figure 36...................................................................... 17 Added Output Filtering Section ................................................... 19 Changes to Controller Mode Section........................................... 19 Changes to Response Time Capability Section .......................... 18 Changes to Table 6.......................................................................... 22 Changes to Figure 47, Figure 48, and Figure 49 ......................... 23 Updated Outline Dimensions ....................................................... 24 Changes to Ordering Guide .......................................................... 24 7/04--Rev. 0: Initial Version
Rev. A | Page 2 of 24
AD8318 SPECIFICATIONS
VPOS = 5 V, CLPF = 220 pF, TA = 25C, 52.3 termination resistor at INHI, unless otherwise noted. Table 1.
Parameter SIGNAL INPUT INTERFACE Specified Frequency Range DC Common-Mode Voltage MEASUREMENT MODE f = 900 MHz Input Impedance 1 dB Dynamic Range Conditions INHI (Pin 14) and INLO (Pin 15) Min 0.001 VPOS - 1.8 VOUT (Pin 6) shorted to VSET (Pin 7), sinusoidal input signal RTADJ = 500 TA = +25C -40C < TA < +85C 1 dB error 1 dB error 957||0.71 57 48 -1 -58 -24.5 22 0.78 1.52 0.0011 0.003 523||0.68 57 50 -2 -59 -24.4 20.4 0.73 1.35 ||pF dB dB dBm dBm mV/dB dBm V V dB/C dB/C ||pF dB dB dBm dBm mV/dB dBm V V Typ Max 8 Unit GHz V
Maximum Input Level Minimum Input Level Slope Intercept Output Voltage--High Power In PIN = -10 dBm Output Voltage--Low Power In PIN = -40 dBm Temperature Sensitivity PIN = -10 dBm 25C TA 85C -40C TA +25C f = 1.9 GHz RTADJ = 500 Input Impedance 1 dB Dynamic Range TA = +25C -40C < TA < +85C Maximum Input Level 1 dB error Minimum Input Level 1 dB error Slope Intercept Output Voltage--High Power PIN = -10 dBm In Output Voltage--Low Power PIN = -35 dBm In Temperature Sensitivity PIN = -10 dBm 25C TA 85C -40C TA +5C f = 2.2 GHz RTADJ = 500 Input Impedance 1 dB Dynamic Range TA = +25C -40C < TA < +85C Maximum Input Level 1 dB error Minimum Input Level 1 dB error Slope Intercept Output Voltage--High Power In PIN = -10 dBm Output Voltage--Low Power In PIN = -35 dBm Temperature Sensitivity PIN = -10 dBm 25C TA 85C -40C TA +25C
-26 19.5 0.7 1.42
-23 24 0.86 1.62
-27 17 0.63 1.2
-22 24 0.83 1.5
0.0011 0.0072 391||0.66 58 50 -2 -60 -24.4 19.6 0.73 1.34 -0.0005 0.0062
dB/C dB/C ||pF dB dB dBm dBm mV/dB dBm V V dB/C dB/C
-28 15 0.63 1.2
-21.5 25 0.84 1.5
Rev. A | Page 3 of 24
AD8318
Parameter f = 3.6 GHz Input Impedance 1 dB Dynamic Range Conditions RTADJ = 51 TA = +25C -40C < TA < +85C 1 dB error 1 dB error Min Typ 119||0.7 58 42 -2 -60 -24.3 19.8 0.717 1.46 0.0022 0.004 33||0.59 57 48 -1 -58 -24.3 25 0.86 1.59 0.0033 0.0069 60 58 3 -55 -23 37 1.06 1.78 0.028 -0.0085 4.9 25 60 60 45 PIN = 2.2 GHz; -10 dBm, fNOISE = 100 kHz, CLPF = 220 pF PIN = Off to -10 dBm, 90% to 10% PIN = -10 dBm to off, 10% to 90% 90 10 12 Max Unit ||pF dB dB dBm dBm mV/dB dBm V V dB/C dB/C ||pF dB dB dBm dBm mV/dB dBm V V dB/C dB/C dB dB dBm dBm mV/dB dBm V V dB/C dB/C V mV mA MHz MHz nV/Hz ns ns
Maximum Input Level Minimum Input Level Slope Intercept Output Voltage--High Power In PIN = -10 dBm Output Voltage--Low Power In PIN = -40 dBm Temperature Sensitivity PIN = -10 dBm 25C TA 85C -40C TA +25C f = 5.8 GHz Input Impedance 1 dB Dynamic Range RTADJ = 1000 TA = +25C -40C < TA < +85C 1 dB error 1 dB error
Maximum Input Level Minimum Input Level Slope Intercept Output Voltage--High Power In PIN = -10 dBm Output Voltage--Low Power In PIN = -40 dBm Temperature Sensitivity PIN = -10 dBm +25C TA +85C -40C TA +25C f = 8.0 GHz 3 dB Dynamic Range RTADJ = 500 TA = +25C -40C < TA < +85C 3 dB error 3 dB error
Maximum Input Level Minimum Input Level Slope Intercept Output Voltage--High Power In PIN = -10 dBm Output Voltage--Low Power In PIN = -40 dBm Temperature Sensitivity PIN = -10 dBm +25C TA +85C -40C TA +25C OUTPUT INTERFACE Voltage Swing Output Current Drive Small Signal Bandwidth Video Bandwidth (or Envelope Bandwidth) Output Noise Fall Time Rise Time VOUT (Pin 6) VSET = 0 V; PIN = -10 dBm, no load 1 VSET = 2.1 V; PIN = -10 dBm, no load1 VSET = 1.5 V; PIN = -50 dBm PIN = -10 dBm; from CLPF to VOUT
Rev. A | Page 4 of 24
AD8318
Parameter VSET INTERFACE Nominal Input Range Logarithmic Scale Factor Bias Current Source TEMPERATURE REFERENCE Output Voltage Temperature Slope Current Source/Sink POWER-DOWN INTERFACE Logic Level to Enable Device ENBL Current When Enabled ENBL Current When Disabled POWER INTERFACE Supply Voltage Quiescent Current vs. Temperature Supply Current when Disabled vs. Temperature
1 2
Conditions VSET (Pin 7) PIN = 0 dBm; measurement mode 2 PIN = -65 dBm; measurement mode2 PIN = -10 dBm; VSET = 2.1 V TEMP (Pin 13) TA = 25C, RLOAD = 10 k -40C TA +85C, RLOAD = 10 k TA = 25C ENBL (Pin 16) ENBL = 5 V ENBL = 0 V; sourcing VPSI (Pin 3 and Pin 4), VPSO (Pin 9) ENBL = 5 V -40C TA +85C ENBL = 0 V, total currents for VPSI and VPSO -40C TA +85C
Min
Typ 0.5 2.1 -0.04 2.5
Max
Unit
V dB/mV A 0.63 V mV/C mA V A A 5.5 82 V mA A/C A A
0.57
0.6 2 10/0.1 1.7 <1 15
4.5 50
5 68 150 260 350
Controller mode. Gain = 1. For other gains, see the Measurement Mode section.
Rev. A | Page 5 of 24
AD8318 ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Supply Voltage: Pin VPSO, Pin VPSI ENBL, VSET Voltage Input Power (Single-Ended, re: 50 ) Internal Power Dissipation JA1 Maximum Junction Temperature Operating Temperature Range Storage Temperature Range Lead Temperature
1
Rating 5.7 V 0 to VPOS 12 dBm 0.73 W 55C/W 125C -40C to +85C -65C to +150C 260C
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
With package die paddle soldered to thermal pads with vias connecting to inner and bottom layers.
ESD CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily accumulate on the human body and test equipment and can discharge without detection. Although this product features proprietary ESD protection circuitry, permanent damage may occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD precautions are recommended to avoid performance degradation or loss of functionality.
Rev. A | Page 6 of 24
AD8318 PIN CONFIGURATION AND FUNCTION DESCRIPTIONS
12 11 10 9
CMIP CMIP TADJ VPSO
13 14 15 16
TEMP INHI
CMOP 8 VSET 7
AD8318
INLO ENBL CMIP CMIP
1 2
VOUT 6 CLPF 5 VPSI
3
VPSI
4
Figure 2. Pin Configuration
Table 3. Pin Function Descriptions
Pin No. 1, 2, 11, 12 3, 4 5 6 7 8 9 10 13 14 15 16 Mnemonic CMIP VPSI CLPF VOUT VSET CMOP VPSO TADJ TEMP INHI INLO ENBL Paddle Description Device Common (Input System Ground). Positive Supply Voltage (Input System): 4.5 V to 5.5 V. Voltage on Pin 3, Pin 4, and Pin 9 should be equal. Loop Filter Capacitor. Measurement and Controller Output. Setpoint Input for Controller Mode or Feedback Input for Measurement Mode. Device Common (Output System Ground). Positive Supply Voltage (Output System): 4.5 V to 5.5 V. Voltage on Pin 3, Pin 4, and Pin 9 should be equal. Temperature Compensation Adjustment. Temperature Sensor Output. RF Input. Nominal input range: -60 dBm to 0 dBm (re: 50 ), ac-coupled. RF Common for INHI. AC-coupled RF common. Device Enable. Connect to VPSI for normal operation. Connect pin to ground for disable mode. Internally Connected to CMIP (Solder to Ground).
Rev. A | Page 7 of 24
04853-002
AD8318 TYPICAL PERFORMANCE CHARACTERISTICS
VPOS = 5 V, TA = +25C, -40C, +85C; CLPF = 220 pF; RTADJ = 500 ; unless otherwise noted. Colors: +25C
2.2 2.0 1.8 1.6 1.4
VOUT (V)
Black; -40C
Blue; +85C
Red.
2.0 1.6 1.2 0.8
ERROR (dB)
04853-008
2.0 1.6 1.2 0.8
ERROR (dB)
2.2 2.0 1.8 1.6 1.4
VOUT (V)
0.4 0 -0.4 -0.8 -1.2
0.4 0 -0.4 -0.8 -1.2 -1.6 -55 -45 -35 -25 -15 PIN (dBm) -5 5 -2.0 15
1.2 1.0 0.8 0.6 0.4 0.2 -65 -55 -45 -35 -25 -15 PIN (dBm) -5 5
1.2 1.0 0.8 0.6
04853-003
-1.6 -2.0 15
0.4 0.2 -65
Figure 3. VOUT and Log Conformance vs. Input Amplitude at 900 MHz, Typical Device
2.2 2.0 1.8 1.6 1.4
VOUT (V)
Figure 6. VOUT and Log Conformance vs. Input Amplitude at 3.6 GHz, Typical Device, RTADJ = 51
2.2 2.0 1.8 1.6 2.0 1.6 1.2 0.8
ERROR (dB) ERROR (dB)
04853-007
2.0 1.6 1.2 0.8
ERROR (dB)
0.4 0 -0.4 -0.8 -1.2 -1.6 -55 -45 -35 -25 -15 PIN (dBm) -5 5 -2.0 15
1.4
VOUT (V)
0.4 0 -0.4 -0.8 -1.2 -1.6 -55 -45 -35 -25 -15 PIN (dBm) -5 5 -2.0 15
1.2 1.0 0.8 0.6 0.4 0.2 -65
1.2 1.0 0.8 0.6
04853-004
0.4 0.2 -65
Figure 4. VOUT and Log Conformance vs. Input Amplitude at 1.9 GHz, Typical Device
2.2 2.0 1.8 1.6 1.4
VOUT (V)
Figure 7. VOUT and Log Conformance vs. Input Amplitude at 5.8 GHz, Typical Device, RTADJ = 1000
2.2 2.0 1.8 1.6 4.5 3.6 2.7 1.8 0.9 0 -0.9 -1.8 -2.7 -3.6 -4.5 -55 -45 -35 -25 PIN (dBm) -15 -5 5
2.0 1.6 1.2 0.8
ERROR (dB)
0.4 0 -0.4 -0.8 -1.2 -1.6 -55 -45 -35 -25 -15 PIN (dBm) -5 5 -2.0 15
1.4
VOUT (V)
1.2 1.0 0.8 0.6 0.4 0.2 -65
1.2 1.0 0.8 0.6
04853-005
0.4 0.2 -65
Figure 5. VOUT and Log Conformance vs. Input Amplitude at 2.2 GHz, Typical Device
Figure 8. VOUT and Log Conformance vs. Input Amplitude at 8 GHz, Typical Device
Rev. A | Page 8 of 24
04853-006
AD8318
2.0 1.6 1.2 0.8
ERROR (dB)
ERROR (dB)
04853-009
2.0 1.6 1.2 0.8 0.4 0 -0.4 -0.8 -1.2 -1.6 -2.0 -65 -55 -45 -35 -25 -15 PIN (dBm) -5 5 15
04853-012
0.4 0 -0.4 -0.8 -1.2 -1.6 -2.0 -65 -55 -45 -35 -25 -15 PIN (dBm) -5 5 15
Figure 9. Distribution of Error over Temperature After Ambient Normalization vs. Input Amplitude at 900 MHz for at Least 70 Devices
Figure 12. Distribution of Error at Temperature After Ambient Normalization vs. Input Amplitude at 3.6 GHz for at Least 70 Devices, RTADJ = 51
2.0 1.6 1.2 0.8
2.0 1.6 1.2 0.8
ERROR (dB)
0.4 0 -0.4 -0.8 -1.2
04853-010
ERROR (dB)
0.4 0 -0.4 -0.8 -1.2 -1.6 -2.0 -65 -55 -45 -35 -25 -15 PIN (dBm) -5 5 15
04853-013
-1.6 -2.0 -65 -55 -45 -35 -25 -15 PIN (dBm) -5 5 15
Figure 10. Distribution of Error at Temperature After Ambient Normalization vs. Input Amplitude at 1900 MHz for at Least 70 Devices
2.0 1.6 1.2 0.8
Figure 13. Distribution of Error at Temperature After Ambient Normalization vs. Input Amplitude at 5.8 GHz for at Least 70 Devices, RTADJ = 1000
4.5 3.6 2.7 1.9
ERROR (dB)
0 -0.4 -0.8 -1.2
04853-011
ERROR (dB)
0.4
0.9 0 -0.9 -1.8 -2.7 -3.6 -4.5 -65 -55 -45 -35 -25 PIN (dBm) -15 -5 5
04853-014
-1.6 -2.0 -65 -55 -45 -35 -25 -15 PIN (dBm) -5 5 15
Figure 11. Distribution of Error at Temperature After Ambient Normalization vs. Input Amplitude at 2.2 GHz for at Least 70 Devices
Figure 14. Distribution of Error at Temperature After Ambient Normalization vs. Input Amplitude at 8 GHz for at Least 70 Devices
Rev. A | Page 9 of 24
AD8318
j1 j2 j0.5
10k RF OFF
j0.2
NOISE SPECTRAL DENSITY (nV/ Hz)
1k -40dBm -20dBm -60dBm
0
0.2 8GHz
0.5
1
2 0.1GHz
5.8GHz -j0.2
0.9GHz 1.9GHz 3.6GHz
100 -10dBm
04853-018
2.2GHz -j2
04853-015
0dBm
-j0.5 START FREQUENCY = 0.1GHz STOP FREQUENCY = 8GHz
10 1 3 10 30 100 300 FREQUENCY (kHz) 1k 3k 10k
-j1
Figure 15. Input Impedance vs. Frequency; No Termination Resistor on INHI, ZO = 50
0.07 0.06
Figure 18. Noise Spectral Density of Output; CLPF = Open
1k
SUPPLY CURRENT (A)
0.05
0.04 0.03
DECREASING VENBL
INCREASING VENBL
NOISE SPECTRAL DENSITY (nV/ Hz)
100
0.02
04853-016
0.01 0 1.4
1.5
1.6 VENBL (V)
1.7
1.8
10 1 3 10 30 100 300 FREQUENCY (kHz) 1k 3k 10k
Figure 16. Supply Current vs. Enable Voltage
Figure 19. Noise Spectral Density of Output Buffer (from CLPF to VOUT); CLPF = 0.1 F
2.2 2.0 2.0 1.6 1.2 0.8
VOUT
1.8 1.6
1.2 1.0 0.8 0.6 0.4
0 -0.4 -0.8 -1.2 -1.6 -55 -45 -35 -25 -15 PIN (dBm) -5 5 -2.0 15
PULSED RF INPUT 0.1GHz, -10dBm GND
04853-017
0.2 -65
20ns PER HORIZONTAL DIVISION
Figure 17. VOUT Pulse Response Time; Pulsed RF Input 0.1 GHz, -10 dBm; CLPF = Open
Figure 20. Output Voltage Stability vs. Supply Voltage at 1.9 GHz When VP Varies by 10%, Multiple Devices
Rev. A | Page 10 of 24
04853-020
ERROR (dB)
200mV/VERTICAL DIVISION
VOUT (V)
1.4
0.4
04853-019
AD8318 THEORY OF OPERATION
The AD8318 is a 9-stage demodulating logarithmic amplifier that provides RF measurement and power amplifier control functions. The design of the AD8318 is similar to the AD8313 logarithmic detector/controller. However, the AD8318 input frequency range extends to 8 GHz with 60 dB dynamic range. Other improvements include: reduced intercept variability vs. temperature, increased dynamic range at higher frequencies, low noise measurement and controller output (VOUT), adjustable low-pass corner frequency (CLPF), temperature sensor output (TEMP), negative transfer function slope for higher accuracy, and 10 ns response time for RF burst detection capability. A block diagram is shown in Figure 21.
VPSI ENBL TADJ VPSO
CMIP, the input system common pin, provides a quality low impedance connection to the printed circuit board (PCB) ground via four package pins. Ground the package paddle, which is internally connected to the CMIP pin, to the PCB to reduce thermal impedance from the die to the PCB. The logarithmic function is approximated in a piecewise fashion by nine cascaded gain stages. For a more complete explanation of the logarithm approximation, refer to the AD8307 data sheet, available at www.analog.com. The cells have a nominal voltage gain of 8.7 dB each and a 3 dB bandwidth of 10.5 GHz. Using precision biasing, the gain is stabilized over temperature and supply variations. Since the cascaded gain stages are dc-coupled, the overall dc gain is high. An offset compensation loop is included to correct for offsets within the cascaded cells. At the output of each of the gain stages, a square-law detector cell rectifies the signal. The RF signal voltages are converted to a fluctuating differential current with an average value that increases with signal level. Along with the nine gain stages and detector cells, an additional detector is included at the input of the AD8318, altogether providing a 60 dB dynamic range. After the detector currents are summed and filtered, the function ID x log10(VIN/VINTERCEPT) is formed at the summing node, where: ID is the internally set detector current. VIN is the input signal voltage. VINTERCEPT is the intercept voltage (that is, when VIN = VINTERCEPT, the output voltage would be 0 V if capable of going to 0 V).
TEMP
TEMP SENSOR
GAIN BIAS
SLOPE
I
V
VSET
I DET INHI INLO DET DET DET
V
VOUT
CLPF
CMIP
CMOP
Figure 21. Block Diagram
A fully differential design, using a proprietary high speed SiGe process, extends high frequency performance. Input INHI receives the signal with a low frequency impedance of nominally 1200 in parallel with 0.7 pF. The maximum input with 1 dB log conformance error is typically 0 dBm (re: 50 ). The noise spectral density referred to the input is 1.15 nV/Hz, which is equivalent to a voltage of 118 V rms in a 10.5 GHz bandwidth, or a noise power of -66 dBm (re: 50 ). This noise spectral density sets the lower limit of the dynamic range. However, the low end accuracy of the AD8318 is enhanced by specially shaping the demodulating transfer characteristic to partially compensate for errors due to internal noise.
04853-021
Rev. A | Page 11 of 24
AD8318 USING THE AD8318
BASIC CONNECTIONS
The AD8318 is specified for operation up to 8 GHz. As a result, low impedance supply pins with adequate isolation between functions are essential. In the AD8318, VPSI and VPSO, the two positive supply pins, must be connected to the same positive potential. The VPSI pin biases the input circuitry, while VPSO biases the low noise output driver for VOUT. Separate commons are also included in the device. CMOP is used as the common for the output drivers. Pins CMIP and CMOP should be connected to a low impedance ground plane. A power supply voltage of between 4.5 V and 5.5 V should be applied to VPSO and VPSI. In addition, 100 pF and 0.1 F power supply decoupling capacitors connect close to each power supply pin. The two adjacent VPSI pins can share a pair of decoupling capacitors due to their proximity.
1
VPSI 40k DISCHARGE ENBL 200 2 x VBE
ENABLE CMIP
Figure 23. ENBL Interface
: 2.07V @: 2.07V
VS 499 (SEE TEXT) C5 0.1F
CH1 500mV
12 11 10 9
04853-051
C6 100pF
M400ns A CH1 T 425.200ns
920mV
CMIP CMIP TADJ VPSO TEMP OUT RF INPUT
13 TEMP C1 1nF 14 INHI C2 1nF 15 INLO 16 ENBL
Figure 24. ENBL Response Time; VPOS = 5.0 V; Input AC-Coupling Caps = 18 pF; CLPF = Open
CMOP 8 VSET 7
R1 52.3
AD8318
VOUT 6 CLPF 5 VPSI
3
INPUT SIGNAL COUPLING
VOUT (SEE TEXT)
CMIP CMIP VS
1 2
VPSI
4
C7 100pF C8 0.1F VS
Figure 22. Basic Connections
The paddle of the AD8318 LFCSP package is internally connected to CMIP. For optimum thermal and electrical performance, solder the paddle to a low impedance ground plane.
The RF input to the AD8318 (INHI) is single-ended and must be ac-coupled. INLO (input common) should be ac-coupled to ground (see Figure 22). Suggested coupling capacitors are 1 nF ceramic 0402 style capacitors for input frequencies of 1 MHz to 8 GHz. The coupling capacitors should be mounted close to the INHI and INLO pins. These capacitor values can be increased to lower the input stage high-pass cutoff frequency. The highpass corner is set by the input coupling capacitors and the internal 10 pF capacitor. The dc voltage on INHI and INLO is approximately one diode voltage drop below the voltage applied to the VPSI pin. The Smith chart in Figure 15 shows the AD8318 input impedance vs. frequency. Table 4 lists the reflection coefficient and impedance at select frequencies. For Figure 15 and Table 4, the 52.3 input termination resistor is removed. At dc, the resistance is typically 2 k. At frequencies up to 1 GHz, the impedance is approximated as 1000 ||0.7 pF. The RF input pins are coupled to a network as shown in the simplified schematic in Figure 25.
ENABLE INTERFACE
To enable the AD8318, the ENBL pin must be pulled high. Taking ENBL low puts the AD8318 in sleep mode, reducing current consumption to 260 A at ambient. The voltage on ENBL must be greater than 2 VBE (~1.7 V) to enable the device. When enabled, the ENBL pin draws less than 1 A. When ENBL is pulled low, the pin sources 15 A. The enable interface has high input impedance. An internal 200 resistor is placed in series with the ENBL input for added protection. Figure 23 depicts a simplified schematic of the enable interface. The response time of the AD8318 ENBL interface is shown in Figure 24.
04853-022
Rev. A | Page 12 of 24
04853-023
40k
2 x VBE
AD8318
VPSO
VPSI 10pF 10pF FIRST GAIN STAGE 2k INLO Gm STAGE OFFSET COMP
04853-024
CURRENT
CLPF 10 + 0.2V - VOUT
20k INHI
20k
150 200
04853-025
A = 8.6dB
CMOP
Figure 26. Output Interface
SETPOINT INTERFACE
The setpoint interface is shown in Figure 27. The VSET input drives the high impedance (250 k) input of an internal operational amplifier. The VSET voltage appears across the internal 3.13 k resistor to generate ISET. When a portion of VOUT is applied to VSET, the feedback loop forces -ID x log10 (VIN/VINTERCEPT) = ISET. If VSET = VOUT/X, Then ISET = VOUT/(X x 3.13 k). The result is VOUT = (-ID x 3.13 k x X) x log10(VIN/VINTERCEPT).
ISET VSET
Figure 25. Input Interface
While the input can be reactively matched, this is typically not necessary. An external 52.3 shunt resistor (connected on the signal side of the input coupling capacitors, see Figure 22) combines with the relatively high input impedance to provide an adequate broadband 50 match. Table 4. Input Impedance for Select Frequency
Frequency MHz 100 450 900 1900 2200 3600 5300 5800 8000 S11 Real Imaginary 0.918 -0.041 0.905 -0.183 0.834 -0.350 0.605 -0.595 0.524 -0.616 0.070 -0.601 -0.369 -0.305 -0.326 -0.286 -0.390 -0.062 Impedance (Series) 927-j491 173-j430 61-j233 28-j117 28-j102 26-j49 20-j16 22-j16 22-j3
The coupling time constant, 50 x CC/2, forms a high-pass corner with a 3 dB attenuation at fHP = 1/(2 x 50 x CC ), where C1 = C2 = CC. Using the typical value of 1 nF, this high-pass corner is ~3.2 MHz. In high frequency applications, fHP should be as large as possible to minimize the coupling of unwanted low frequency signals. Likewise, in low frequency applications, a simple RC network forming a low-pass filter should be added. This should generally be placed at the generator side of the coupling capacitors, thereby lowering the required capacitance value for a given highpass corner frequency.
3.13k CMOP
04853-026
Figure 27. VSET Interface
The slope is given by -ID x X x 3.13 k = -500 mV x X. For example, if a resistor divider to ground is used to generate a VSET voltage of VOUT/2, then X = 2. The slope is set to -1 V/decade or -50 mV/dB.
OUTPUT INTERFACE
The logarithmic output interface is shown in Figure 26. The VOUT pin is driven by a PNP output stage. An internal 10 resistor is placed in series with the emitter follower output and the VOUT pin. The rise time of the output is limited mainly by the slew on CLPF. The fall time is an RC limited slew provided by the load capacitance and the pull-down resistance at VOUT. There is an internal pulldown resistor of 350 . Any resistive load at VOUT is placed in parallel with the internal pull-down resistor and provides additional discharge current.
TEMPERATURE COMPENSATION OF OUTPUT VOLTAGE
The AD8318 functionality includes the capability to externally trim the temperature drift. Attaching a ground-referenced resistor to the TADJ pin alters an internal current, minimizing intercept drift vs. temperature. As a result, the Resistor RTADJ can be optimized for operation at different frequencies.
ICOMP 2V
VINTERNAL ~0.4V 2k TADJ
04853-027
Figure 28. TADJ Interface
Rev. A | Page 13 of 24
AD8318
RTADJ, nominally 500 for optimal temperature compensation at 2.2 GHz input frequency, is connected between the TADJ pin and ground (see Figure 22). The value of this resistor partially determines the magnitude of an analog correction coefficient that is employed to reduce intercept drift. Table 5 lists recommended resistors for various frequencies. These resistors provide the best overall temperature drift based on measurements of a diverse population of devices. The relationship between output temperature drift and frequency is nonlinear and is not easily modeled. Experimentation is required to choose the correct RTADJ resistor at frequencies not listed in Table 5. Table 5. Recommended RTADJ Resistors
Frequency 900 MHz 1.9 MHz 2.2 GHz 3.6 GHz 5.8 GHz 8 GHz Recommended RTADJ 500 500 500 51 1 k 500
MEASUREMENT MODE
When the VOUT voltage, or a portion of the VOUT voltage, is fed back to VSET, the device operates in measurement mode. As shown in Figure 30, the AD8318 has an offset voltage, a negative slope, and a VOUT measurement intercept greater than its input signal range.
2.4 2.1 1.8 1.5 VOUT 25C ERROR 25C 2.0 1.5 1.0
1.2 0.9 0.6 0.3 0 -65 -60 -55 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 PIN (dBm) RANGE OF CALCULATION OF SLOPE AND INTERCEPT
0 -0.5 -1.0
0
5
10 15 INTERCEPT
Figure 30. Typical Output Voltage vs. Input Signal
TEMPERATURE SENSOR
The AD8318 internally generates a voltage that is proportional to absolute-temperature (VPTAT). The VPTAT voltage is multiplied by a factor of 5, resulting in a 2 mV/C output at the TEMP pin. The output voltage at 27C is typically 600 mV. An emitter follower drives the TEMP pin, as shown in Figure 29.
VPSI INTERNAL TEMP 4k 1k CMIP
The output voltage vs. input signal voltage of the AD8318 is linear-in-dB over a multidecade range. The equation for this function is VOUT = X x VSLOPE/DEC x log10(VIN/VINTERCEPT) = X x VSLOPE/dB x 20 x log10(VIN/VINTERCEPT) (1) (2)
where: X is the feedback factor in VSET = VOUT/X. VINTERCEPT is expressed in Vrms. VSLOPE/DEC is nominally -500 mV/decade or -25 mV/dB. VINTERCEPT, expressed in dBV, is the x-axis intercept of the linearin-dB transfer function shown in Figure 30. VINTERCEPT is 7 dBV (20 dBm, re: 50 or 2.239 Vrms) for a sinusoidal input signal. The slope of the transfer function can be increased to accommodate various converter mV per dB (LSB per dB) requirements. However, increasing the slope can reduce the dynamic range. This is due to the limitation of the minimum and maximum output voltages, determined by the chosen scaling factor X. The minimum value for VOUT is X x VOFFSET. The offset voltage, VOFFSET, is equal to 0.5 V and is internally added to the detector output signal. VOUT(MIN) = (X x VOFFSET)
Figure 29. Temp Sensor Interface
The internal pull-down resistance is 5 k. The temperature sensor has a slope of 2 mV/C. The temperature sensor output varies with output current due to increased die temperature. Output loads less than 1 k draw enough current from the output stage causing this increase to occur. An output current of 10 mA results in the voltage on the temperature sensor to increase by 1.5C, or ~3 mV. Best precision from the temperature sensor is obtained when the supply current to AD8318 remains fairly constant, that is, no heavy load drive.
04853-028
Rev. A | Page 14 of 24
04853-029
-1.5
ERROR (dB)
0.5
VOUT (V)
AD8318
The maximum output voltage is 2.1 V x X, and cannot exceed 400 mV below the positive supply. VOUT(MAX) = (2.1 V x X) when X < (VPOS - 400 mV)/(2.1 V) VOUT(MAX) = (VPOS - 400 mV) when X (VPOS - 400 mV)/(2.1 V) When X = 1, the typical output voltage swing is 0.5 V to 2.1 V. The output voltage swing is modeled using the equations above and restricted by the following equation: VOUT(MIN) < VOUT < VOUT(MAX) When X = 4 and VPOS = 5 V (X x VOFFSET) < VOUT < (VPOS - 400 mV) (4 x 0.5 V) < VOUT < (2.1 V x 4) 2 V < VOUT < 4.6 V For X = 4, Slope = -100 mV/dB; VOUT can swing 2.6 V, and the usable dynamic range is reduced to 26 dB from 0 dBm to -26 dBm. The slope is very stable vs. process and temperature variation. When base-10 logarithms are used, VSLOPE/DECADE represents the output voltage per decade of input power. One decade is equal to 20 dB; VSLOPE/DECADE/20 = VSLOPE/dB represents the output voltage slope in volts/dB. As noted in the previous equations, the VOUT voltage has a negative slope. This is the correct slope polarity to control the gain of many power amplifiers and other VGAs in a negative feedback configuration. Since both the slope and intercept vary slightly with frequency, refer to Table 1 for application-specific values for slope and intercept. Although demodulating log amps respond to input signal voltage, not input signal power, it is customary to discuss the amplitude of high frequency signals in terms of power. In this case, the characteristic impedance of the system, Z0, must be known to convert voltages to corresponding power levels. Beginning with the definitions of dBm and dBV, P(dBm) = 10 x log10(Vrms2/(Z0 x 1 mW)) V(dBV) = 20 x log10(Vrms/1 Vrms) When Equation 3 is expanded P(dBm) = 20 x log10(Vrms) - 10 x log10(Z0 x 1 mW) and given Equation 4, Equation 5 can be rewritten as P(dBm) = V(dBV) - 10 x log10(Z0 x 1 mW) (6) (5) (3) (4)
VOUT2 1.6 1.4 1.0 2.2 2.0 1.8
DEVICE CALIBRATION AND ERROR CALCULATION
The measured transfer function of the AD8318 at 2.2 GHz is shown in Figure 31. The figure shows plots of both output voltage vs. input power and calculated log conformance error vs. input power. As the input power varies from -65 dBm to 0 dBm, the output voltage varies from 2 V to about 0.5 V.
VOUTIDEAL = SLOPE x (PIN - INTERCEPT) SLOPE = (VOUT1 - VOUT2)/(PIN1 - PIN2) INTERCEPT = PIN1 - (VOUT1/SLOPE) ERROR (dB) = (VOUT x VOUTIDEAL)/SLOPE VOUT +25C VOUT -40C VOUT +85C ERROR +25C ERROR -40C ERROR +85C
2.5 2.0 1.5
1.2 1.0 0.8 0.6 0.4 0.2 -65 -60 -55 -45 -40 -35 -30 -25 -20 -15 PIN1 PIN2 PIN (dBm) -5 0
0 -0.5 -1.0 -1.5 -2.0
VOUT1
5 INTERCEPT
Figure 31. Transfer Function at 2.2 GHz
Because slope and intercept vary from device to device, boardlevel calibration is performed to achieve high accuracy. The equation can be rewritten for output voltage, from the Measurement Mode section, using an intercept expressed in dBm VOUT = Slope x (PIN - Intercept) (8)
In general, the calibration is performed by applying two known signal levels to the AD8318 input and measuring the corresponding output voltages. The calibration points are generally chosen to be within the linear-in-dB operating range of the device (see Figure 31). Calculation of slope and intercept is done using the equations Slope = (VOUT1 - VOUT2)/(PIN1 - PIN2) Intercept = PIN1 - VOUT1/Slope (9) (10)
Once slope and intercept are calculated, an equation can be written to allow calculation of an (unknown) input power based on the output voltage of the detector. PIN (unknown) = VOUT (measured)/Slope + Intercept (11)
For example, PINTERCEPT for a sinusoidal input signal, expressed in terms of dBm (decibels referred to 1 mW), in a 50 system is PINTERCEPT(dBm) = VINTERCEPT (dBV) - 10 x log10(Z0 x 1 mW) = +7 dBV - 10 x log10(50 x 10 ) = +20 dBm For further information on the intercept variation dependence upon waveform, refer to the AD8313 and AD8307 data sheets.
Rev. A | Page 15 of 24
-3
(7)
04853-030
ERROR (dB)
0.5
VOUT (V)
AD8318
Using the equation for the ideal output voltage (7) as a reference, the log conformance error of the measured data can be calculated Error(dB) = (VOUT(MEASURED) - VOUT(IDEAL))/Slope (12) Figure 33 shows how calibration points can be adjusted to increase dynamic range, but at the expense of linearity. In this case, the calibration points for slope and intercept are set at -4 dBm and -60 dBm. These points are at the end of the linear range of the device. Once again, at 25C, an error of 0 dB is seen at the calibration points. Note also that the range over which the AD8318 maintains an error of < 1 dB is extended to 60 dB at 25C and 58 dB over temperature. The disadvantage of this approach is that linearity suffers, especially at the top end of the input range.
2.2 2.0 1.8 1.6 VOUT +25C VOUT -40C VOUT +85C ERROR +25C ERROR -40C ERROR +85C 2.5 2.0 1.5 1.0
Figure 31 includes a plot of the error at 25C, the temperature at which the log amp is calibrated. Note that the error is not zero. This is because the log amp does not perfectly follow the ideal VOUT vs. PIN equation, even within its operating region. The error at the calibration points (-12 dBm and -52 dBm, in this case) is, however, equal to zero by definition. Figure 31 includes error plots for the output voltage at -40C and +85C. These error plots are calculated using the slope and intercept at 25C. This method is consistent with a mass-production environment where calibration at temperature is not practical.
1.2 1.0 0.8 0.6 58dB DYNAMIC RANGE (1dB ERROR) 0.4 0.2 -65 -60 -55 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 PIN (dBm) 0 5
0 -0.5 -1.0 -1.5 -2.0 -2.5
In some applications, very high accuracy is required at just one power level or over a reduced input range. For example, in a wireless transmitter, the accuracy of the high power amplifier (HPA) is most critical at, or close to, full power. Figure 32 shows the same measured data as Figure 31. Note that accuracy is very high from -10 dBm to -30 dBm. Below -30 dBm, the error increases to about -1 dB. This is because the calibration points have changed to -14 dBm and -26 dBm.
VOUT +25C VOUT -40C VOUT +85C ERROR +25C ERROR -40C ERROR +85C 2.5 2.0 1.5 1.0
Figure 33. Dynamic Range Extension by Choosing Calibration Points Close to the End of the Linear Range
2.2 2.0 1.8 1.6 1.4
1.2 1.0
0 -0.5 -1.0
ERROR (dB)
0.5
Another way of presenting the error function of a log amp detector is shown in Figure 34. In this case, the dB error at hot and cold temperatures is calculated with respect to the output voltage at ambient. This is a key difference in comparison to the plots in Figure 32 and Figure 33. Previously, all errors were calculated with respect to the ideal transfer function at ambient. When this alternative technique is used, the error at ambient becomes, by definition, equal to 0 (see Figure 34). This is valid if the device transfer function perfectly follows the ideal VOUT = Slope x (Pin - Intercept) equation. However, because a log amp in practice never perfectly follows this equation (especially outside of its linear operating range), this plot tends to artificially improve linearity and extend the dynamic range. This plot is a useful tool for estimating temperature drift at a particular power level with respect to the (nonideal) output voltage at ambient. However, to achieve this level of accuracy in an end application requires calibration at multiple points in the operating range of the device.
VOUT2
VOUT1 0.6 0.4 0.2 -65 -60 -55 -50 -45 -40 -35 -30 PIN (dBm) PIN2 -20 -10 -5 PIN1 0 5
VOUT (V)
-1.5 -2.0 -2.5
04853-031
Figure 32. Output Voltage and Error vs. PIN with 2-Point Calibration at -10 dBm and -30 dBm
Calibration points are chosen to suit the application at hand. In general, the calibration points are never chosen in the nonlinear portion of the transfer function of the log amp (above -5 dBm or below -60 dBm, in this case).
Rev. A | Page 16 of 24
04853-038
ERROR (dB)
VOUT (V)
SELECTING CALIBRATION POINTS TO IMPROVE ACCURACY OVER A REDUCED RANGE
1.4
0.5
AD8318
2.2 2.0 1.8 1.6 1.4 VOUT +25C VOUT -40C VOUT +85C ERROR +25C wrt VOUT ERROR -40C wrt VOUT ERROR +85C wrt VOUT 2.5 2.0 1.5 1.0
reduced temperature range, higher measurement accuracy is achievable for all frequencies.
VOUT +25C VOUT 0C ERROR -10C ERROR +70C VOUT -40C VOUT +70C ERROR -20C VOUT -10C ERROR -40C VOUT +85C ERROR +25C ERROR 0C VOUT -20C ERROR +85C
ERROR (dB)
0.5 0 -0.5 -1.0 -1.5 -2.0 -2.5 0 5
VOUT (V)
2.2 2.0 1.8 1.6 1.4
2.5 2.0 1.5 1.0
1.2 1.0 0.8 0.6 0.4 0.2 -65 -60 -55 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 PIN (dBm)
1.2 1.0 0.8 0.6 0.4 0.2 -65 -60 -55 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 PIN (dBm) 0 5
0 -0.5 -1.0 -1.5 -2.0 -2.5
Figure 34. Error vs. Temperature with Respect to Output Voltage at 25C (Does Not Take Transfer Function Nonlinearities at 25C into Account)
VARIATION IN TEMPERATURE DRIFT FROM DEVICE TO DEVICE
Figure 35 shows a plot of output voltage and error for multiple AD8318 devices measured at 5.8 GHz. The concentration of black error plots represents the performance of the population at 25C (slope and intercept are calculated for each device). The red and blue curves indicate the measured behavior of a population of devices over temperature. This suggests a range on the drift (from device to device) of 1.2 dB.
2.2 2.0 1.8 1.6 1.4 2.0 1.6 1.2 0.8
Figure 36. Typical Drift at 5.8 GHz for Various Temperatures
SETTING THE OUTPUT SLOPE IN MEASUREMENT MODE
To operate in measurement mode, VOUT is connected to VSET. This yields the typical logarithmic slope of -25 mV/dB. The output swing corresponding to the specified input range is then approximately 0.5 V to 2.1 V. The slope and output swing can be increased by placing a resistor divider between VOUT and VSET (that is, one resistor from VOUT to VSET and one resistor from VSET to common). As an example, if two equal resistors, such as 10 k/10 k, are used, the slope doubles to approximately -50 mV/dB. The input impedance of VSET is approximately 500 k. Slope setting resistors should be kept below ~50 k to prevent this input impedance from affecting the resulting slope. When increasing the slope, the new output voltage range cannot exceed the output voltage swing capability of the output stage. Refer to the Measurement Mode section for further details.
AD8318
VOUT 10k VSET
04853-033
1.2 1.0 0.8 0.6 0.4 0.2 -65 -55 -45 -35 -25 -15 PIN (dBm) -5 5
0 -0.4 -0.8 -1.2 -1.6 -2.0 15
04853-050
ERROR (dB)
0.4
VOUT (V)
Figure 35. Output Voltage and Error vs. Temperature (+25C, -40C, and +85C) of a Population of Devices Measured at 5.8 GHz
-50mV/dB
TEMPERATURE DRIFT AT DIFFERENT TEMPERATURES
Figure 36 shows the log slope and error over temperature for a 5.8 GHz input signal. Error due to drift over temperature consistently remains within 0.5 dB, and only begins to exceed this limit when the ambient temperature drops below -20C. When using a
10k
Figure 37. Increasing the Slope
Rev. A | Page 17 of 24
04853-039
ERROR (dB)
0.5
VOUT (V)
04853-032
AD8318
+5V PULSED RF INPUT 1nF INHI 52.3 1nF INLO GND 50 50 -5.2V -5.2V 100 VPOS VOUT VSET AD8318 OUTPUT +5V 50
40
AD8318
ADCMP563
50 100
04853-040
COMPARATOR OUTPUT
VREF = 1.8V-1.2V
Figure 38. AD8318 Operating with the High Speed ADCMP563 Comparator
RESPONSE TIME CAPABILITY
The AD8318 has a 10 ns rise/fall time capability (10% to 90%) for input power switching between the noise floor and 0 dBm. This capability enables RF burst measurements at repetition rates beyond 45 MHz. In most measurement applications, the AD8318 has an external capacitor connected to CLPF to provide additional filtering for VOUT. However, using the CLPF capacitor slows the response time as does stray capacitance on VOUT. For an application requiring maximum RF burst detection capability, the CLPF pin is left unconnected. In this case, the integration function is provided by the 1.5 pF on-chip capacitor. There is a 10 internal resistor in series with the output driver. Because of this resistor, it is necessary to add an external 40 backterminating resistor in series with the output when driving a 50 load. Place the back-terminating resistor close to the VOUT pin. The AD8318 has the drive capability to drive a 50 load at the end of a coaxial cable or transmission line when back terminated (see Figure 38). The circuit diagram in Figure 38 shows the AD8318 used with a high speed comparator circuit. The 40 series resistor at the output of the AD8318 combines with an internal 10 to properly match to the 50 input of the comparator.
PULSED RF INPUT -50dB -30dB -20dB -10dB
Figure 39 shows the response of the AD8318 and the comparator for a 500 MHz pulsed sine wave of varying amplitudes. The output level of the AD8318 is the signal strength of the input signal. For applications where these RF bursts are very small, the output level does not change by a large amount. Using a comparator is beneficial in this case because it turns the output of the log amp into a limiter-like signal. While this configuration does result in the loss of received signal power level, it does allow for presence-only detection of low power RF bursts.
OUTPUT FILTERING
For applications in which maximum video bandwidth and, consequently, fast rise time are desired, it is essential that the CLPF pin be left unconnected and free of any stray capacitance. To reduce the nominal output video bandwidth of 45 MHz, connect a ground-referenced capacitor (CFLT) to the CLPF pin, as shown in Figure 40. Generally, this is done to reduce output ripple (at twice the input frequency for a symmetric input waveform, such as sinusoidal signals).
AD8318
ILOG VOUT +4 3.13k 1.5pF CLPF
04853-042
CFLT
Figure 40. Lowering the Postdemodulation Bandwidth
AD8318 OUTPUT
CFLT is selected using the following equation:
COMPARATOR OUTPUT
C FLT =
04853-041
( x 3.13 k x VideoBandw idth )
1
- 1.5 pF (13)
0
100
200
300
400
500
600
700
800
TIME (ns)
Figure 39. Pulse Response of AD8318 and Comparator for RF Pulses of Varying Amplitudes
Set the video bandwidth to a frequency equal to about onetenth the minimum input frequency. This ensures that the output ripple of the demodulated log output, which is at twice the input frequency, is well filtered.
Rev. A | Page 18 of 24
AD8318
In many log amp applications, it may be necessary to lower the corner frequency of the postdemodulation filtering to achieve low output ripple while maintaining a rapid response time to changes in signal level. For an example of a 4-pole active filter, see the AD8307 data sheet. The basic connections for operating the AD8318 as an analog controller with the AD8367 are shown in Figure 42. The AD8367 is a low frequency to 500 MHz VGA with 45 dB of dynamic range. This configuration is very similar to the one shown in Figure 41. For applications working at high input frequencies, such as cellular bands or WLAN, or those requiring large gain control ranges, the AD8318 can control the ADL5330 10 MHz to 3 GHz RF VGA. For further details and an application schematic, refer to the ADL5330 data sheet. The voltage applied to the GAIN pin controls the gain of the AD8367. This voltage, VGAIN, is scaled linear-in-dB with a slope of 20 mV/dB and runs from 50 mV at -2.5 dB of gain, up to 1.0 V at +42.5 dB. The incoming RF signal to the AD8367 has a varying amplitude level. Receiving and demodulating it with the lowest possible error requires that the signal levels be optimized for the highest signal-to-noise ratio (SNR) feeding into the analog-to-digital converters (ADC). This is done by using an automatic gain control (AGC) loop. In Figure 42, the voltage output of the AD8318 modifies the gain of the AD8367 until the incoming RF signal produces an output voltage that is equal to the setpoint voltage VSET.
+3V RF INPUT SIGNAL VPOS GND INPT GAIN 0.1F VOUT 57.6 HPFL CHP 100pF RHP 100 RF OUTPUT SIGNAL 174
CONTROLLER MODE
The AD8318 provides a controller mode feature at the VOUT pin. Using VSET for the setpoint voltage, it is possible for the AD8318 to control subsystems, such as power amplifiers (PAs), variable gain amplifiers (VGAs), or variable voltage attenuators (VVAs) that have output power that increases monotonically with respect to their gain control signal. To operate in controller mode, the link between VSET and VOUT is broken. A setpoint voltage is applied to the VSET input; VOUT is connected to the gain control terminal of the VGA and the detector RF input is connected to the output of the VGA (usually using a directional coupler and some additional attenuation). Based on the defined relationship between VOUT and the RF input signal when the device is in measurement mode, the AD8318 adjusts the voltage on VOUT (VOUT is now an error amplifier output) until the level at the RF input corresponds to the applied VSET. When the AD8318 operates in controller mode, there is no defined relationship between VSET and VOUT voltage; VOUT settles to a value that results in the correct input signal level appearing at INHI/INLO. In order for this output power control loop to be stable, a groundreferenced capacitor is connected to the CLPF pin. This capacitor, CFLT, integrates the error signal (in the form of a current) to set the loop bandwidth and ensure loop stability. For further details on control loop dynamics, refer to the AD8315 data sheet.
AD8367
VGA
R2 261
R1 1k
+5V
100MHz BANDPASS FILTER
DAC
+VSET SETPOINT VOLTAGE
VOUT VSET
VPOS INHI INLO
1nF
AD8318
CLPF
04853-047
VGA/VVA DIRECTIONAL COUPLER ATTENUATOR 1nF INHI 52.3 GAIN CONTROL VOLTAGE VOUT
RFIN
CFLT 100pF
GND
1nF
Figure 42. AD8318 Operating in Controller Mode to Provide Automatic Gain Control Functionality in Combination with the AD8367
AD8318
INLO 1nF VSET
DAC
CLPF CFLT
04853-034
The AGC loop is capable of controlling signals over ~45 dB dynamic range. The output of the AD8367 is designed to drive loads 200 . As a result, it is not necessary to use the 53.6 resistor at the input of the AD8318; the nominal input impedance of 2 k is sufficient. If the AD8367 output drives a 50 load, such as an oscilloscope or spectrum analyzer, use a simple resistive divider network. The divider used in Figure 42 has an insertion loss of 11.5 dB. Figure 43 shows the transfer function of output power vs. VSET voltage for a 100 MHz sine wave at -40 dBm into the AD8367.
Figure 41. AD8318 Controller Mode
Decreasing VSET, which corresponds to demanding a higher signal from the VGA, tends to increase VOUT. The gain control voltage of the VGA must have a positive sense. A positive control voltage to the VGA increases the gain of the device.
Rev. A | Page 19 of 24
AD8318
0 -5 -10 -15 -20 1.2 1.0 0.8 0.6 0.4
10 0 MAXIMUM INPUT LEVEL -10 -20
ERROR (dB)
POUT (dBm)
PIN (dBm)
-25 -30 -35 -40 -45 -50 -55 -60 0.6 0.8 1.0 1.2 1.4 VSET (V) 1.6 1.8
0.2 0 -0.2 -0.4 -0.6 -0.8
-30 -40 MINIMUM INPUT LEVEL -50 -60
04853-048
-1.0 -1.2 2.0
-70 -80 0.5
0.6
0.7
0.8
0.9
1.0 1.2 VSET (V)
1.2
1.3
1.4
1.5
Figure 43. AD8367 Output Power vs. AD8318 Setpoint Voltage
Figure 45. Setpoint Voltage vs. Input Power. Optimal signal levels must be used to achieve the full 45 dB dynamic range capabilities of the AD8367.
For the AGC loop to remain locked, the AD8318 must track the envelope of the VGA output signal and provide the necessary voltage levels to the AD8367 gain control input. Figure 44 shows an oscilloscope screen image of the AGC loop depicted in Figure 42. A 50 MHz sine wave with 50% AM modulation is applied to the AD8367. The output signal from the VGA is a constant envelope sine wave with an amplitude corresponding to a setpoint voltage at the AD8318 of 1.0 V.
AM MODULATED INPUT
In some cases, if VGAIN is >1.0 V it can take an unusually long time for the AGC loop to recover; that is, the output of the AD8318 remains at an abnormally high value and the gain is set to its maximum level. A voltage divider is placed between the output of the AD8318 and the AD8367 GAIN pin to ensure that VGAIN does not exceed 1.0 V. In Figure 42, CHP and RHP are configured to reduce oscillation and distortion due to harmonics at higher gain settings. Some additional filtering is recommended between the output of the AD8367 and the input of the AD8318. This helps to decrease the output noise of the AD8367, which can reduce the dynamic range of the loop at higher gain settings (smaller VSET). Response time and the amount of signal integration are controlled by CFLT. This functionality is analogous to the feedback capacitor around an integrating amplifier. Though it is possible to use large capacitors for CFLT, in most applications, values under 1 nF provide sufficient filtering.
1
AD8318 V OUT
2
AD8367 OUTPUT
3
04853-045
CH1 50.0mV CH3 20.0mV
CH2 200mV
M4.00ms
A CH2
64.0mV
Figure 44. Oscilloscope Screen Image Showing an AM Modulated Input Signal to the AD8367. The AD8318 tracks the envelope of this input signal and applies the appropriate voltage to ensure a constant output from the AD8367.
Calibration in controller mode is similar to the method used in measurement mode. Do a simple two-point calibration by applying two known VSET voltages or DAC codes and measuring the output power from the VGA. Slope and intercept are calculated with the following equations: Slope = (VSET1 - VSET2)/(POUT1 - POUT2) Intercept = POUT1 - VSET1/Slope VSET = Slope x (Px - Intercept) For more information on AGC applications, refer to the AD8367 data sheet or ADL5330 data sheet. (14) (15) (16)
The 45 dB control range is constant for the range of VSET voltages. The input power levels to the AD8367 must be optimized to achieve this range. In Figure 45, the minimum and maximum input power levels are shown vs. setpoint voltage.
Rev. A | Page 20 of 24
04853-049
AD8318
CHARACTERIZATION SETUP AND METHODS
The general hardware configuration used for the AD8318 characterization is shown in Figure 46. The primary setup used for characterization is measurement mode. The characterization board is similar to the customer evaluation board with the exception that the RF input has a Rosenberger SMA connector and R10 has changed to a 1 k resistor to remove cable capacitance from the bench characterization setup. Slope and intercept are calculated in this data sheet and in the production environment using linear regression from -50 dBm to -10 dBm. The slope and intercept generate an ideal line. Log conformance error is the difference from the ideal line and the measured output voltage for a given temperature in dB. For additional information on the error calculation, refer to the Device Calibration and Error Calculation section. The hardware configuration for pulse response measurement replaces the 0 series resistor at the VOUT pin with a 40 resistor; the CLPF pin remains open. Pulse response time is measured using a Tektronix TDS51504 Digital Phosphor Oscilloscope. Both channels on the scope are configured for 50 termination. The 10 internal series resistance at VOUT, combined with the 40 resistor, attenuates the output voltage level by two. RF input frequency is set to 100 MHz with -10 dBm at the input of the device. The RF burst is generated using a Rohde & Schwarz SMT06 with the pulse option with a period of 1.5 s, a width of 0.1 s, and a pulse delay of 0.04 s. The output response is triggered using the video output from the SMT06. Refer to Figure 46 for an overview of the test setup.
R AND S SMT06
VIDEO OUT RF OUT -7dBm 3dB SPLITTER 1nF 52.3 1nF 5V
TEKTRONIX TDS51504
CH1* CH3* TRIGGER *50 TERMINATION
VPOS INHI VOUT
40
AD8318
INLO GND
04853-046
VSET
Figure 46. Pulse Response Measurement Test Setup
To measure noise spectral density, the 0 resistor in series with the VOUT pin is replaced with a 1 F dc blocking capacitor. The capacitor is used because the Rohde & Schwarz FSEA spectrum analyzer cannot handle dc voltages at its RF input. The CLPF pin is left open for data collected for Figure 18. For Figure 19, a 1 F capacitor is placed between CLPF and ground. The large capacitor filters the noise from the detector stages of the log amp. Noise spectral density measurements are taken using the FSEA and the SMT06 signal generator. The signal generator frequency is set to 2.2 GHz. The spectrum analyzer has a span of 10 Hz, resolution bandwidth of 50 Hz, video bandwidth of 50 Hz, and averages the signal 100 times. Data is adjusted to account for the dc blocking capacitor impedance on the output at lower frequencies.
Rev. A | Page 21 of 24
AD8318 EVALUATION BOARD
Table 6. Evaluation Board (Rev. A) Bill of Materials
Component VP, GND SW1, R3 Function Supply and Ground Connections Device Enable: When in Position A, the ENBL pin is connected to VP and the AD8318 is in operating mode. In Position B, the ENBL pin is grounded through R3, putting the device in power-down mode. The ENBL pin may be exercised by a pulse generator connected to ENBL SMA and SW1 in Position B. Input Interface: The 52.3 resistor (R1) combines with the AD8318 internal input impedance to give a broadband input impedance of 50 . C1 and C2 are dc-blocking capacitors. A reactive impedance match can be implemented by replacing R1 with an inductor and C1 and C2 with appropriately-valued capacitors. Temperature Sensor Interface: The temperature sensor output voltage is available at the SMA labeled TEMP, via the current limiting resistor, R2. Temperature Compensation Interface: The internal temperature compensation resistor is optimized for an input signal of 2.2 GHz when R4 is 500 . This circuit can be adjusted to optimize performance for other input frequencies by changing the value of Resistor R4. See the Temperature Compensation of Output Voltage section. Output Interface--Measurement Mode: In measurement mode, a portion of the output voltage is fed back to the VSET pin via R7. The magnitude of the slope at VOUT can be increased by reducing the portion of VOUT that is fed back to VSET. R10 can be used as a backterminating resistor or as part of a single-pole low-pass filter. Output Interface--Controller Mode: In this mode, R7 must be open. In controller mode, the AD8318 can control the gain of an external component. A setpoint voltage is applied to the VSET pin, the value of which corresponds to the desired RF input signal level applied to the AD8318 RF input. The magnitude of the control voltage is optionally attenuated, via the voltage divider comprised of R8 and R9, or a capacitor can be installed in R8 to form a low-pass filter along with R9. See the Controller Mode section for more details. Power Supply Decoupling: The nominal supply decoupling consists of a 100 pF filter capacitor placed physically close to the AD8318, a 0 series resistor, and a 0.1 F capacitor placed closer to the power supply input pin. Default Conditions Not Applicable SW1 = A R3 = 10 (Size 0603)
R1, C1, C2
R1 = 52.3 (Size 0402) C1 = 1 nF (Size 0402) C2 = 1 nF (Size 0402) R2 = 1 (Size 0402)
R2
R4
R4 = 499 (Size 0603)
R7, R8, R9, R10
R7 = 0 = (Size 0402) R8 = open (Size 0402) R9 = open (Size 0402 R10= 0 (Size 0402) R7 = open (Size 0402) R8 = open (Size 0402) R9 = 0 (Size 0402) R10 = 0 (Size 0402)
R7, R8, R9, R10
C5, C6, C7, C8, R5, R6
C9
Loop Filter Capacitor: The low-pass corner frequency of the circuit that drives the VOUT pin can be lowered by placing a capacitor between CLPF and ground. Increasing this capacitor increases the overall rise/fall time of the AD8318 for pulsed input signals. See the Output Filtering section for more details.
C5 = 0.1 F (Size 0603) C6 = 100 pF (Size 0402) C7 = 100 pF (Size 0402) C8 = 0.1 F (Size 0603) R5 = 0 (Size 0603) R6 = 0 (Size 0603) C9 = open (Size 0603)
Rev. A | Page 22 of 24
AD8318
VPOS R4 499 (SEE TEXT) C5 0.1F C6 100pF
R5 0
10 9
12
11
R2 1k TEMP C1 1nF RFIN R1 52.3 C2 1nF
CMIP CMIP TADJ VPSO
13 TEMP 14 INHI
CMOP 8 VSET 7
R8 OPEN R7 0 R9 OPEN R10 0 C9 OPEN (SEE TEXT) VSET
AD8318
15 INLO 16 ENBL
VOUT 6 CLPF 5 VPSI
3
VOUT
A ENBL R3 10k
B
CMIP CMIP VPOS
1 2
VPSI
4
SW1 R6 0 C7 100pF
GND
VP
VPOS
Figure 47. Evaluation Board Schematic
04853-035
C8 0.1F
04853-036
Figure 48. Component Side Layout
Figure 49. Component Side Silkscreen
Rev. A | Page 23 of 24
04853-037
AD8318 OUTLINE DIMENSIONS
4.00 BSC SQ 0.60 MAX 0.60 MAX
13 12 16 1
PIN 1 INDICATOR 2.25 2.10 SQ 1.95 0.25 MIN 1.95 BSC
PIN 1 INDICATOR
TOP VIEW
0.65 BSC 3.75 BSC SQ 0.75 0.60 0.50
EXPOSED PAD
(BOTTOM VIEW)
9 8 5
4
12 MAX 1.00 0.85 0.80
0.80 MAX 0.65 TYP 0.05 MAX 0.02 NOM
SEATING PLANE
0.30 0.23 0.18
0.20 REF
COPLANARITY 0.08
COMPLIANT TO JEDEC STANDARDS MO-220-VGGC
Figure 50. 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ] 4 mm x 4 mm Body, Very Thin Quad (CP-16-4) Dimensions shown in millimeters
ORDERING GUIDE
Model AD8318ACPZ-REEL7 1 AD8318ACPZ-WP1, 2 AD8318-EVAL
1 2
Temperature Range -40C to +85C -40C to +85C
Package Description 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ] 16-Lead Lead Frame Chip Scale Package [LFCSP_VQ] Evaluation Board
Package Option CP-16-4 CP-16-4
Ordering Quantity 1500 64
Z = Pb-free part. WP = waffle pack.
(c)2006 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D04853-0-1/06(A)
Rev. A | Page 24 of 24

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